Reconfigurable hybrid beamforming MIMO receiver with inter-band carrier aggregation and RF-domain LMS weight adaptation

ABSTRACT

A reconfigurable, multi-band hybrid beamforming architecture is introduced. The present invention is related to a Cartesian-Combining architecture to efficiently implement RF beamforming for a single downconversion chain employing direct downconversion in which the Cartesian-Combining architecture is extended to hybrid beamforming and to heterodyne downconversion.

RELATED APPLICATIONS

This application is a divisional of U.S. patent application Ser. No.16/163,374, filed Oct. 17, 2018, which claims the benefit of U.S.Provisional Patent Application No. 62/707,024, filed Oct. 17, 2017. Theentire contents of these applications are incorporated herein byreference.

GOVERNMENT INTEREST

This invention was made with government support under contractECCS-1343324 awarded by the National Science Foundation. The governmenthas certain rights in the invention.

BACKGROUND OF THE INVENTION

Directional communication using electronically steered antenna arrayswill be a key ingredient in future wireless networks in the traditionalsub-6 GHz bands and in the new millimeter-wave frequency bands. Infirst-generation mm-wave networks (e.g. IEEE 802.11ad wireless LAN),phased-array beamforming is used to obtain a directional and steerableantenna pattern. Due to its simplicity and power efficiency, the RFbeamformer architecture shown in FIG. 1. View (A) has emerged as thepreferred approach to perform the algorithmically simple spatial signalprocessing required in a phased array, namely to apply programmablecomplex-valued weights to the signals received at the elements of anantenna array. Application of the complex-valued weights can beimplemented in the RF-domain through one of several approaches includingphase-shifter/variable-gain amplifier combinations, or through vectormodulators as shown in view (B). Another approach is to use theCartesian-combining architecture shown in view (C).

Advanced multi-antenna-based spatial signal processing techniques arenecessary to achieve higher spectral efficiency, network capacity andbetter interference management in future millimeter-wave networks. Thedigital beamformer architecture shown in FIG. 2 offers the highestflexibility in implementing such spatio-temporal signal processing.However, the high power consumption of the local oscillator (LO)distribution network, data converters and digital signal processingmakes digital beamforming infeasible for a large number of antennaelements. Hybrid beamformers seek to strike a compromise by performingthe bulk of the spatial processing for a large number of antennas at RF,along with a handful of downconversion chains to facilitate digitalspatio-temporal processing. There are two types of hybridbeamformers—the “partially-connected or sub-array” type (i.e., PC-HBF)of FIG. 3 and the “fully-connected” type (i.e., FC-HBF) of FIG. 4. Thefully-connected type can offer superior performance when compared to thepartially-connected type at the expense of greater implementationcomplexity. The partially-connected type can be implemented usingexisting RF-domain phased arrays such as the one shown in FIG. 1 whileimplementations of the fully-connected type are not previously known inthe literature.

It is anticipated that future millimeter-wave networks will be deployedin several widely separated bands. So far, the 28, 37, 39, 45, 57-71,71-76, 81-86, and 94 GHz bands have been identified for commercial use.Deployment of standards will initially be in a few of the lowerfrequency bands, but the use of increasingly higher frequency bands willbe necessary to address the anticipated demand for capacity and datarates. Another likely scenario is the adoption of different frequencybands in different regions. Therefore, we anticipate the need forreconfigurable, flexible beamformers that can operate in areconfigurable manner in many widely separated frequency bands.

SUMMARY OF THE INVENTION

Herein is described a reconfigurable, multi-band, fully-connected hybridbeamforming architecture. The architecture has N_(R) inputs(corresponding to N_(R) antennas) with No downconversion chains. Thesignal received at each of the N_(R) antennas is optionally amplified bya low-noise amplifier (LNA).

Complex-weights, one for every downconversion chain, may be applied tothe output of each LNA, or directly to the signal received from theantennas. The complex weights may be applied using one of threealternative approaches. In a first embodiment, a cascade of aphase-shifter and a variable-gain amplifier may be used. In a second,alternative embodiment, the complex weights may be applied using avector modulator. In a third, alternative embodiment, one pair ofprogrammable gain amplifiers per downconversion chain may be used toapply the complex weights. Following complex weighting using any of theabove embodiments, the signals from the individual chains from eachantenna are combined (i.e., summed), and then applied to the inputs ofthe N_(D) downconversion chains.

The weighted signals can be combined in different ways, for example, bysumming currents or by summing voltages or by using power combinercircuits. Furthermore, the combiner can be split into multiple stages,and a different combining topology can be used in each stage.

The circuitry preceding the first mixer stage is designed to support abandwidth spanning (f_(L), f_(H)). Each downconversion chain mayincorporate an independent local oscillator (LO) generation circuit, asshown in FIG. 5. The architecture of the present invention allowscomplete flexibility in the selection of frequencies and streams in eachdownconversion chain. For example, if the LO generation circuits areconfigured to generate and distribute N_(D) distinct LO frequencies tothe first stage mixers in the N_(D) chains, the architecture behaveslike N_(D) independent phased arrays at each frequency between f_(L) andf_(H). As another example, if all LO generation circuits are configuredto produce the same frequency, then N_(D) distinct streams can besupported by the N_(D) downconversion chains.

Non-image-reject embodiments hold for the case when a single LOgeneration circuit is shared between all N_(D) downconversion chains.Each of the N_(D) downconversion chains can be reconfigured into threenon-image-reject modes, NIR1 NIR2 and NIR3. In NIR1, simple(non-quadrature) downconversion is performed in the first stage mixerpair, and full (complex-valued) downconversion is performed in thesecond stage. The advantage of this configuration is that quadraturegeneration of the high frequency first local oscillator is notnecessary, thereby simplifying its generation and distribution. Thedisadvantage of NIR1 is that the beamforming operation is not completeduntil the output of the second stage. In other words, no spatialfiltering is achieved prior to the input of the second stage. Therefore,interferers arriving from directions other than the main lobe directionare not attenuated until the output of the second stage whichnecessitates high-linearity mixers in the first as well as the secondstages.

In the NIR2 mode, signals are weighted by the real and imaginary pathamplifier, summed with a quadrature first LO and then combined in thefirst stage. This is followed by quadrature downconversion in the secondmixing stage.

In the NIR3 mode, signals are weighted by the real and imaginary pathamplifier. A complex-quadrature downconversion is performed in the firstmixing stage, followed by simple (non-quadrature) downconversion in thesecond mixing stage.

The pros and cons of modes NIR2 and NIR3 are the opposite of NIR1.Precise quadrature generation is required in the first stage.Imperfections in quadrature generation lead to vector combining withnon-90° vector combining, which leads to errors in the complex-weightingfunction. However, since the beamforming is completed at the output ofthe first stage, any interferers are spatially filtered prior to thesecond stage mixing quad, thereby easing its linearity requirements.

Each downconverter can be configured into one of two image-reject modesIR1 or IR2, as shown in FIG. 6, using a single first-stage LO frequencyf_(LO1). In this mode, one of two desired frequency channels f_(RF,HI)and f_(RF,LO) in two distinct frequency bands can be received while theother is rejected. To do so, an LO with frequency f_(LO1) is chosen suchthat f_(LO1)=(f_(RF,HI)+f_(RF,LO))/2. By configuring the sign of thesummation at the combiner at the output of the mixing stages, eitherf_(RF,HI) or f_(RF,LO) can be received and the other rejected.

A subset of downconverters (N_(U) in number) among the total N_(D)downconverters can be configured into an lower-sideband image-rejectmode and the remaining N_(L) (=N_(D)−N_(U)) downconverters can beconfigured into a upper-sideband image-reject mode to downconvert onlyf_(RF,HI) or f_(RF,LO), respectively (where f_(RF,HI) and f_(RF,LO)denote two desired frequency channels in two distinct bands).

The image-reject configurations achieve perfect image rejection onlywith ideal quadrature LO signals (i.e., LO signals of equal amplitudeand exactly 90° phase difference). In practice, both LO1 and LO2 willhave quadrature errors. Calibration circuitry is necessary to separatelycorrect quadrature errors in each mixing stage.

A quadrature error correction scheme is described herein which detectsthe quadrature error in the first stage using a pair of mixers, thenuses a digital control loop to actuate a correction mechanism inside theRF-domain weighting amplifiers to correct for quadrature phase errors.Quadrature amplitude errors are eliminated by using limiting amplifiersbetween the LO generating circuit and the mixer.

Two methods are described to adaptively adjust the beamformingcomplex-weights to optimize the beam pattern under a minimum mean-squareerror (MMSE) constraint. Techniques to achieve this goal are well-knownfor digital beamformers but the lack of access to sampled/basebandsignals from each individual antenna makes these techniques infeasible.The two techniques are called Double-Sampling Time-Multiplexed LMS(DS-TM-LMS) and Multi-Stream Time-Multiplexed LMS (MS-TM-LMS), andenable dynamic MMSE adaptation in hybrid beamformers.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a general schematic diagram of a prior art RF beamformerarchitecture.

FIG. 2 is a general schematic diagram of a prior art digital beamformingreceiver.

FIG. 3 is a general schematic diagram of a prior art partially-connectedhybrid beamforming receiver.

FIG. 4 is a schematic diagram of a prior art simplified receiverimplementing the fully-connected hybrid beam former architecture thatcombines complex-weighted signals from antennas prior to downconversionand digitization through multiple downconversion chains. The concept isillustrated here for two downconversion chains but can be generalized toany number.

FIG. 5 is a block schematic diagram of a simplified architecture of afully-connected hybrid beamforming receiver in accordance with thepresent invention, showing the Cartesian-combining architecture toperform independent complex-weighting from each antenna element to eachof several downconversion chains, followed by direct downconversion ineach downconversion chain.

FIG. 6, view (A) is a block schematic diagram of a simplifiedarchitecture of a concurrent dual-band beamforming receiver inaccordance with the present invention, showing, in view (B),image-reject complex-mixing based Cartesian complex-weighting, allowingconcurrent dual-band RF phase shifting.

FIG. 7 shows several different architectures for non-image-reject (NIR)heterodyne Cartesian phase shifting with vector modulator in view (A),complex quadrature mixing in second stage—NIR1 in view (B), complexmixing in first stage and quadrature mixing in second stage—NIR2 in view(C), and complex quadrature mixing in first stage—NIR3 in view (D).

FIG. 8 shows several different architectures for image-reject (IR)Cartesian phase shifting with vector modulator followed by a “Hartley”receiver in view (A) and complex quadrature mixing in both stages inview (B).

FIG. 9 is a block schematic diagram showing an exemplary dual-bandCartesian combining fully-connected hybrid beamforming receiverfunctional front-end structure and a detailed schematic of afour-element two-stream multi-mode adaptive hybrid beamformer prototypehaving stream #1 highlighted in blue and stream 2 highlighted in red.

FIG. 10 shows the LO₁ quadrature error extraction, calibration andcorrection (at LO-path) and LO₂ quadrature error correction at BB.

FIG. 11, view (A) is a block schematic diagram of a DS-TM-LMSadaptation.

FIG. 11, view (B) is a block schematic diagram of a MS-DM-LMSadaptation.

FIG. 12 shows timing diagrams for the DS-TM-LMS in view (A) andMS-TM-LDS adaptations in view (B).

DETAILED DESCRIPTION

Presented herein is a fully-connected hybrid beamforming mm-wavemultiple-input, multiple-output (MIMO) receiver with two keyinnovations. The invention is explained by describing a four antenna,two stream implementation using 28/37 GHz. It should be realized thatthis embodiment is exemplary only, and that the scope of the inventionis meant to cover implementations using any number of antennas producingmultiple streams at multiple frequencies.

First, the receiver can be configured into three modes: two single-bandmulti-stream modes at 28 or 37 GHz that can support single- ormulti-user MIMO, and a concurrent 28/37 GHz dual-band single-streamphased-array inter-band carrier-aggregation mode. In all modes, thereceiver features full-connectivity from each antenna element input toeach output stream, thereby maximizing usage of the available aperture.Second, the digitally programmable RF beamforming weights can becontrolled by an external serial interface, or by an on-chipmixed-signal adaptation loop that implements one of two possible“time-multiplexed” least-mean square (LMS) algorithms—thedouble-sampling time-multiplexed LMS (DS-TM-LMS) or the multi-streamtime-multiplexed LMS (MS-TM-LMS).

Unlike conventional LMS-type adaptation algorithms that require accessto the individual array inputs and the combined output and are thereforenot easily amenable to a hybrid beamformer, both algorithms update theRF-domain weights by accessing only the combined downconverted arrayoutputs. Such adaptation is valuable for adaptive main-lobe, side-lobeor null steering, but more importantly, it can assist or augmentcodebook-based beam acquisition/tracking algorithms, which may fail inthe presence of multipath, on- or off-channel interferers.

A simplified architecture of the four-element, two-stream RF beamformingreceiver of the present invention is shown in FIG. 6. This embodiment,although showing an embodiment using IR mode, can be reconfigured, asdescribed below, to operate in any one of the described NIR modes. Ineach element, a concurrent dual-band (or, alternatively, a wideband) LNAis shared between the two streams. Each stream comprises 28/37 GHzdual-band per-element, per-stream complex-weights, and signal combiners.This is followed by two image-reject downconverters (one per stream)which select either the lower or the higher band using high-side orlow-side LO injection, respectively. While the frequencies of thedesired signals in the two bands can, in general, be chosen to be atsome offset from their image locations, independent LO's would berequired for each downconverter, which adds complexity. Here, thedesired signals in the two bands are chosen to be at the image frequencyof each other. This allows the LO generation circuitry to be sharedbetween the downconversion chains, which facilitates inter-band carrieraggregation without hardware overhead. It is important to note that anyinterferer can be attenuated by spatial filtering or null steering. Forinterference at the image location, this image-reject architectureenables additional suppression. The Cartesian-combining technique iswell-suited to implement programmable RF-domain complex weighting at twowidely separated frequencies. The complex weights are set by the gainratio of a pair of programmable-gain amplifiers (PGA) in conjunctionwith complex-quadrature downconversion, and therefore do not requirefrequency-selective elements unlike conventional phase-shifters. In FIG.6, Cartesian complex weighting is applied at the output of the firstmixing stage, while the cascade of the two mixing stages enablesimage-rejection.

The functional schematic of FIG. 6 shows the proposed fully-connectedhybrid beamformer (FC-HBF) receiver that supports independentcomplex-weighting from 28 GHz and/or 37 GHz band received at eachantenna to each of two heterodyne downconversion chains. There are threemajor considerations in the implementation of this architecture. First,the front-end must have a suitable frequency response to support therequired bands. A contiguously wideband circuit that covers the widelyseparated bands under consideration can be designed but posessignificant challenges. An alternative approach is to design thefront-end circuit to have a “dual-band” frequency response in the twobands under consideration. Second, a generalized application of theCartesian-combining architecture for multiple streams may employ directdownconversion with independent LO generation circuitry in eachdownconversion chain. While direct downconversion may advantageous insome respects, the LO would need to cover a very wide frequency rangespanning the two widely separated bands. Generation and distribution ofthe LO in such an architecture is a challenging task.

Alternatively, a heterodyne version of the Cartesian-combiningarchitecture is advantageous since the required LO range is smaller. Thedisadvantage is that the image-frequency interference must be suppressedthrough combination of appropriate frequency planning and choice ofmixing architecture. Third, RF-domain complex-valued beamforming weightsmust be applied, and the weighted signals combined at two widelyseparated frequencies. Current phased arrays employ different types ofphase shifters which typically have relatively limited bandwidth, inaddition to their other shortcomings. The aforementioned challenges areaddressed in this design using several techniques which are summarizednext.

Through appropriate choice of coupling coefficient, coupled resonatorloads can be used in the front-end circuits including the LNA, PGA's andthe combiner to design for either a dual-band or a contiguously widebandfrequency response, as described above.

A dual-band heterodyne architecture with image-reject downconversionstages is introduced. The LO frequency is chosen such that the desiredsignal bands are located at their mutual image frequency locations. Theimage-reject mixers can be configured to reject the low-side (orhigh-side)-band in each stream with the same LO. This allows seamlessreconfiguration between concurrent dual-band CA mode, where onedownconversion path is configured to reject high-side-band (37 GHz) andother to reject low-side-band (28 GHz), and two multi-stream MIMO modes,where both the downconversion paths are configured to rejecthigh-/low-side-band. Inset of FIG. 10 shows the signal spectrum atdifferent parts of the multi-mode HBF in carrier aggregation (CA) mode.It can be seen that image-band interferer gets rejected using twomechanisms—by front-end null-steering, and by image-rejectdownconversion.

A dual band RF-beamforming technique is introduced. The aforementionedarchitecture can be generalized and developed into several differentvariations. The proposed receiver can be reconfigured amongst all thevariations including an image-reject variant to support different modes.

Complex Weighting and Downconversion Principle

This technique has its roots in a Cartesian combining technique whichuses a pair of programmable-gain amplifiers (PGA) and acomplex-quadrature direct downconversion mixer to perform complexweighting (for RF beamforming) and RF-to-baseband conversion. Thisprinciple was then elegantly extended to a single-stream beamformingreceiver by invoking signal path linearity to sum the complex weightedsignals from each antenna element before complex-quadraturedownconversion, thereby allowing significant simplification in the LOdistribution network. The Cartesian combining technique was extended intwo ways: (1) by introducing a particular approach (shown in view (B) ofFIG. 2) to combine complex-weighting with heterodyne down-conversion,and (2) by introducing a FC-HBF architecture for multi-stream reception.

The evolution of the proposed architectures is described starting withthe structure shown in view (A) of FIG. 7 where complex-weighting isperformed by a 90° combiner (i.e., a quadrature hybrid) and a pair ofPGA's, and is followed by heterodyne downconversion. Typicalimplementations of the 90° combiner have several undesirable propertiesfor on-chip implementation, including high loss, large area, limitedbandwidth and need for matching at the ports. To eliminate an explicit90° combiner, the architectures, shown in views (B)-(D) of FIG. 7 andviews (A)-(C) of FIG. 8 can be used.

Non-Image-Reject (NIR) Architectures

In FIG. 7, view (A), the first downconversion uses a simple mixer, anddoes not offer image rejection. The architectures shown in FIG. 7, views(B)-(D) are derived from the architecture shown in FIG. 7, view (A) andare therefore termed “non-image-reject” (NIR). The NIR1 architectureshown in FIG. 7, view (B) can be derived from FIG. 7, view (A) bytranslating the combiner from RF to baseband, and by absorbing the 90°phase shifter block in the second mixing stage. NIR2, shown in FIG. 7,view (C) can be realized by translating the combining from RF to IF, andby absorbing the 90° phase shifter block in the first mixing stage.NIR3, shown in FIG. 7, view (D), can be realized from NIR1 bycommutating two mixing stages.

These architectures have different advantages and challenges. NIR1 doesnot require high-frequency quadrature LO signals in the first mixingstage; however, since the Cartesian phase shifting operation iscompleted only after the second mixing stage, both mixer stages areexposed to blockers. On the other hand, in both NIR2 and NIR3, Cartesiancomplex-weighting is completed after the first stage of mixing; however,quadrature generation is necessary for the high-frequency LO. ComparingNIR2 and NIR3, NIR2 uses the fewest mixers, but it requires quadratureLO signals in both mixing stages.

Image-Reject (IR) Architectures

In the multi-mode reconfigurable FC-HBF, image rejection is essential inthe CA mode, and desirable to suppress image-frequency interference inthe other two modes. An image-reject architecture can be derivedstarting from the structure of FIG. 8, view (A), which comprises avector modulator followed by a Hartley image-reject receiver. Twotransformations can be performed in each mixing stage as follows: (1)the combining operation that precedes each mixer can be translated afterthe corresponding mixing stage, and (2) each 90° phase shifter can beabsorbed in the subsequent mixing stage, as shown in FIG. 8, view (B).

The image-rejection mechanism of the IR architecture in FIG. 8, view (B)can also be understood mathematically by calculating signals atdifferent nodes of the receive path, labeled A-to-G. To generalize thetreatment to include the NIR modes, the signals in the mixer paths ofFIG. 8, view (B) are multiplied by parameters l₁₋₈, where l=0 signifiesthat the corresponding mixer is turned OFF, and l=+1 or −1 denote thesign of the combining operation that follows the corresponding mixer.The signals at A-to-G can be written as shown in (1) at the bottom ofthe page. Input signals (A) in the high sideband(f_(RFH)=f_(LO1)+f_(LO2)) and low sideband (f_(RFH)=f_(LO1)−f_(LO2)) arerepresented in terms of their complex envelope

(t) and

(t), respectively. The table in FIG. 8, view (C) lists the basebandenvelope for different settings of l₁₋₈.

Multi-Stream Cartesian-Combining FC-HBF

The architectures NIR1-3 and IR can be extended to multiple antennas,resulting in the FC-HBF final receiver architecture of the presentinvention. The resulting structure is referred to herein as theMulti-stream Cartesian Combining FC-HBF. Specifically, to implementcomplex-weighting and image-reject heterodyne downconversion formultiple antennas for a single stream, the structure of FIG. 8, view (B)can be extended by using a pair of PGA's for each antenna in each streamand combining the corresponding PGA outputs. For one stream, thisresults in a structure shown in blue (or red) in FIG. 9. As aconsequence of linearity in the RF signal path, the summation can beimplemented at the input of the first mixing stage. Additional streamscan be supported by replicating the structure depicted in blue (or red).

In an ordinary (i.e., non-Cartesian-combining beamformer) image-rejectreceiver, quadrature error (QE) in both mixing stages can beconsolidated and corrected at BB. However, in a Cartesian-combiningimage-reject receiver, the first stage QE, when captured at BB, varieswith weight settings. To maintain high image-rejection across allcomplex-weight settings, QE from each mixing stage should be calibratedseparately. In the first stage, where significant QE is expected due tothe high frequency and PPF-based quadrature generation, the followingtechnique is used to extract and calibrate QE in LO1 separately.

First, the LO₁ QE is translated to IF (4.5 GHz in measurement) using thetop mixer pair of first mixing stage, as shown in FIG. 10. The QEbetween two mixer outputs at IF is then converted to a voltage using across-coupled mixer pair (a step of ˜20 mV/degree is shown in FIG. 10,as measured from a prototype). Cross-coupled mixers are used to equalizethe loading at two IF outputs, and thus reduce imperfections due to RFand LO trace mismatches inside the OE-extraction circuit. The sign ofthe voltage representing the QE is extracted using a comparator and fedto a digital calibration engine which minimizes the average comparatoroutput by increasing or decreasing the 5-bit control words of thecapacitor banks in tuned-LC I/Q LO buffers, which can tune the I/Qphases with ˜0.75°/LSB phase resolution. This calibration can reduce rawQE of over 20° in a 30-36 GHz LO frequency range to below 1° (See FIG.10). The LO₂ QE is corrected at BB using a phase rotator. In someembodiments, calibration of the 2nd stage only may achieve >35 dB IRRfor a limited number of complex weight settings, while calibration ofboth stages may result in >35 dB over the entire range of weights inboth 28 and 37 GHz bands.

Time-Multiplexed LMS Beam Adaptation Algorithms

Real-time beam pattern adaptation schemes seek to dynamically adaptbeamforming weights under an MMSE criterion. Conventionally, the weightupdate algorithm is expressed in terms ofW(k+1)=w(k)−μ∇_(W)[MSE]  (1)where w is the weight vector, ∇_(w)[MSE] represents the gradient of themean-square error MSE, and μ is the update rate. In general, real-timeestimation of the gradient requires knowledge of the input to eachelement in the beamformer. For example, in the least-mean square (LMS)algorithm, the gradient is estimated by correlating the inputs x withthe error between a “desired” signal d(k) and the beamformer outputw(k)^(H)x(k).w(k+1)=w(k)−μx(k)[d(k)−w(k)^(H) x(k)]^(H)  (2)

Here, d(k) that can be obtained either from a training sequence or fromsymbol decisions during a decision-directed beam-tracking mode or fromsymbol decisions during a decision-directed beam-tracking mode followinginitial training. Implementation of this algorithm is straightforward ina digital beamformer, where all inputs to the beamformer are availablein sampled-data form. However, its implementation in an RF/hybridbeamformer is problematic since the correlation involves sampling an RFsignal (of which only the baseband content is of interest), itsmultiplication with a baseband desired signal, followed by integration.

Two solutions are described. Each is based on two key ideas: (1) The LMSupdates of each weight are independent of other weights and cantherefore be time-multiplexed by calculating one update per cycle. (2)Beamformer input data for a particular element can be extracted using bysetting the complex weight in that path to unity (1e^(j0)) and settingall other weights to zero. This allows access to the baseband content ofthat input alone without requiring extra hardware.

Double-Sampling Time-Multiplexed LMS (DS-TM-LMS)—

This adaptation algorithm, shown in FIG. 11, view (A), is applicable toboth single-stream phased-array, PC-HBF and FC-HBF receivers. It uses anadaptation clock running at twice the frequency of the symbol clock andoperates as follows: (1) During each symbol period, weights are settwice, once at each positive edge of the adaptation clock. Basebandoutputs are sampled twice, once at each negative edge of the adaptationclock, as shown in FIG. 12. (2) In the first half-cycle, the weight ofonly the n^(th) element is set to unity and other weights are set tozero to extract n^(th) element's input (x_(n)). (3) In the secondhalf-cycle, the current set of weights w(k) is applied to all elements,and the output-error-gradient w.r.t. the n^(th) weight is calculatedfrom the beamformer output and the previously extracted n^(th) inputx_(n) (i.e., computing the n^(th) row of matrix equation (2)). (4) Errorgradients with respect to all other weights are sequentially extracted(one per symbol period) in time-multiplexed fashion. (5) At the end ofN^(th) symbol periods (for an N element array) in the k^(th) LMS updatecycle, all the beamforming weights are updated to w(k+1), and the next[(k+1)^(th)] LMS update cycle starts. (6) The adaptation algorithm isterminated at the end of the training preamble in a data packet.

Multi-Stream Time-Multiplexed LMS (MS-TM-LMS)—In an FC-HBF receiver, theavailability of independently weighted downconversion chains from eachantenna can be exploited for adaptation. This results in a secondalgorithm called Multi-Stream Time-Multiplexed LMS (MS-TM-LMS) which isillustrated in FIG. 11, view (B) for an FC-HBF with two chains. Thebeamforming weight of one chain (i.e., chain “A” in view (B) of FIG. 11)is adapted with the help of the other stream (i.e., chain “B” in view(B) of FIG. 11). Note that this is method is not possible in a PC-HBFsince each downconversion chain in a PC-HBF accesses a different subsetof antennas. The MS-TM-LMS algorithm works as follows: (1) In eachsymbol period, weights are set in both chains at the positive edge ofthe adaptation clock and the baseband outputs are sampled from bothstreams at the negative edge of the adaptation clock, shown in FIG. 12.(2) The weights in the auxiliary chain are set to extract n^(th) inputx_(n). Weights in the main chain are set to their current values. Theoutput error is calculated. Then, the output-error gradient with respectto the n^(th) weight is calculated from the output error and x_(n) bothof which are extracted simultaneously. (3) Similar to DS-TM-LMS, errorgradients with respect to all other weights are extracted sequentially;weights are updated once in each N^(th) symbol period.

The availability of additional downconversion chains (N_(D)) in anFC-HBF can be exploited for multi-stream adaptation. Consider an examplewith N_(D)=4 chains. To adapt weights for one, two or three streams,one, two or three chains can be used as the main chains and theremaining three, two or one chains as auxiliary chains. Adaptation canalso be performed for four streams, given four chains, but the DS-TM-LMSalgorithm would have to be used instead.

Comparison Between DS-TM-LMS and MS-TM-LMS

Adaptation in RF/Hybrid BFs: The MS-TM-LMS technique be used only inFC-HBF receivers as more than one chains are necessary to extractbeamformer input and error output simultaneously. On the other hand,DS-TM-LMS can a single-chain RF beamformer, PC- or FC-HBF's.

Beam Tracking: Both algorithms can support beam tracking by using thereceived beamformer output symbol as the “desired” signal for beamadaptation (i.e., decision-directed adaptation). However, in DS-TM-LMS,half the symbol period is used to extract the input signal, which addsperturbation noise into the signal path, potentially degrading the SNR.This is not a problem in MS-TM-LMS.

Adaptation Rate: Adaptation speed of MS-TM-LMS can be increased byincreasing the number of auxiliary chains. For A_(R) auxiliary chains,A_(R) inputs can be simultaneously extracted which results in a speedupby A_(R) times compared to a single auxiliary chain.

Hardware Overhead: DS-TM-LMS does not require extra hardware in the mainsignal path but requires twice the beam switching speed and basebandbandwidth of MS-TM-LMS. In MS-TM-LMS dedicated auxiliary chains arerequired, but only when beam tracking is desired.

To those skilled in the art to which the invention relates, manymodifications and adaptations of the invention will suggest themselves.Implementations provided herein, including sizes, shapes, ratings andspecifications of various components or arrangements of components, anddescriptions of specific manufacturing processes, should be consideredexemplary only and are not meant to limit the invention in any way. Asone of skill in the art would realize, many variations onimplementations discussed herein which fall within the scope of theinvention are possible. Specifically, the invention is meant to includeembodiments using any number of antennas producing multiple streams atmultiple frequencies. Additionally, weighted signals can be combined indifferent ways, as described above. Accordingly, the method andapparatus disclosed herein are not to be taken as limitations on theinvention but as an illustration thereof.

We claim:
 1. A method for weight adaptation in a hybrid beamformingreceiver having N antenna element inputs and M downconversion chains,comprising: receiving N inputs from the N antenna element inputs of thebeamforming receiver using a symbol clock signal running at a firstfrequency and an adaptation clock signal running at a second frequencytwice the first frequency; during each cycle of the symbol clock: a.extracting an n^(th) input from the N inputs of N elements; b. applyinga current set of weights to all N inputs; c. sampling a weighted outputfrom an m^(th) downconversion chain; and d. calculating an errorgradient of a weight corresponding to the n^(th) input; repeating stepsa.-d. in subsequent cycles of the symbol clock until error gradientshave been calculated corresponding to weights for said all N inputs; andupdating the current set of weights based on the calculated errorgradients corresponding to the weights for said all N inputs.
 2. Themethod of claim 1 wherein the N-element beamforming receiver has the Ninputs and one down-converted output.
 3. The method of claim 2 whereinthe n^(th) input is extracted at a first positive edge of the adaptationclock during a current cycle of the symbol clock.
 4. The method of claim3 wherein the n^(th) input is extracted by setting the weight for then^(th) input to unity and all other weights to zero.
 5. The method ofclaim 1 wherein the current set of weights is applied to said all Ninputs at a second positive edge of the adaptation clock during acurrent cycle of the symbol clock.
 6. The method of claim 1 wherein theoutput of the N-element beamformer is sampled at each negative edge ofthe adaptation clock during the current cycle of the symbol clock. 7.The method of claim 1 wherein the error gradient for each n^(th) weightis calculated as an error between a desired signal and an output of thebeamformer.
 8. The method of claim 7 wherein the error gradient for saideach n^(th) weight is calculated based on the equation:W(k+1)=w(k)−μ∇_(W)[MSE] where: w(k+1) is the updated weight (vector);w(k) is the current weight (vector); μ is an update rate; and ∇_(W)[MSE]is the gradient of the mean square error.
 9. The method of claim 7wherein the desired signal is obtained from a training sequence.
 10. Themethod of claim 1 wherein the weights in the set of weights for eachinput are complex weights.
 11. The method of claim 1 wherein the Nelements are part of a single-stream phased-array, partially-connectedor fully-connected hybrid beamforming receiver.
 12. A method for weightadaptation in a multi-stream, fully connected hybrid beamformingreceiver having N antenna elements comprising: receiving N inputs fromthe N antenna elements of the beamforming receiver using a symbol clockand an adaptation clock signal both running at the same frequency, eachof the N-inputs having a at least one main chain and at least oneauxiliary chain; during each cycle of the symbol clock: a. extracting ann^(th) input of the N-inputs; b. calculating an output error of abaseband output of the receiver; c. calculating an error gradient ofweights corresponding to the n^(th) input based on the n^(th) input andthe output error; repeating steps a.-c. in subsequent cycles of thesymbol clock until error gradients have been calculated corresponding toweights for said all N inputs; and updating the current set of weightsfor one or more main streams based on the calculated error gradientscorresponding to the weights for said all N inputs.
 13. The method ofclaim 12 wherein each of the N-inputs has N_(D) chains.
 14. The methodof claim 13 wherein M of the chains in each input are designated as mainchains and N_(D)-M chains are designated as auxiliary chains.
 15. Themethod of claim 12 wherein the n^(th) input is extracted by setting theweight for the n^(th) auxiliary chain to unity and all other weights forothers of the auxiliary chains to zero.
 16. The method of claim 14wherein weights for one of one or more auxiliary streams are set at apositive edge of the adaptation clock during a current cycle of thesymbol clock.
 17. The method of claim 14 wherein weights for the one ormore main streams of the n^(th) input are set at a positive edge of theadaptation clock during a current cycle of the symbol clock.
 18. Themethod of claim 14 wherein the baseband output is sampled at a negativeedge of the adaptation clock during the current cycle of the symbolclock.
 19. The method of claim 12 wherein the error gradients arecalculated as an error between a desired signal and the baseband output.20. The method of claim 19 wherein the desired signal is obtained from atraining sequence.